Hybrid MIMO architecture using lens arrays

ABSTRACT

Various examples are provided related to hybrid multiple-input/multiple-output (MIMO) architectures. Beam steering can be provided using lens arrays. In one example, a hybrid antenna system includes a plurality of lens antenna subarrays (LAS), each of the LAS including a plurality of antenna elements configured to selectively receive a radio frequency (RF) transmission signal from RF processing circuitry, and a lens extending across the plurality of antenna elements. The RF transmission signal can be provided to a selected antenna of the plurality of antenna elements via a switching network and a common phase shifter for transmission. The lens can be configured to steer a RF transmission generated by the selected antenna in a defined direction. The selected antenna can be determined by the switching network configuration.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. utility application entitled,“Hybrid MIMO Architecture Using Lens Arrays,” having Ser. No.16/231,583, filed Dec. 23, 2018, which claims priority to, and thebenefit of, U.S. provisional application entitled “Hybrid MIMOArchitecture Using Lens Arrays” having Ser. No. 62/631,023, filed Feb.15, 2018, all of which are hereby incorporated by reference in theirentireties.

BACKGROUND

Wireless communications within mm-wave bands (28, 38, 60, 73 GHz, andbeyond) attract growing interest due to the diminishing availability ofopen spectrum in lower frequency bands. One of the recognized benefitsof mm-wave communications is the opportunity to employ electricallylarge antenna arrays to overcome the attenuation and blockage challengesin wide area operation. Unfortunately, the cost and power consumption ofmm-wave mixed signal analog/digital components can make it prohibitivelyexpensive to utilize an RF chain and analog-to-digital/digital-to-analogconverter (ADC/DAC) for each of the antenna elements in a large array toenable multiple-input/multiple-output (MIMO) signal processing in thebaseband. Novel architectures capable of simultaneously addressing thechallenges of cost and power have the potential for broad impact in thenext generation of wireless systems.

BRIEF DESCRIPTION OF THE DRAWINGS

Many aspects of the present disclosure can be better understood withreference to the following drawings. The components in the drawings arenot necessarily to scale, emphasis instead being placed upon clearlyillustrating the principles of the present disclosure. Moreover, in thedrawings, like reference numerals designate corresponding partsthroughout the several views.

FIGS. 1A-1C illustrate examples of phased array architectures, inaccordance with various embodiments of the present disclosure.

FIGS. 2A and 2B illustrate examples of lens antenna subarray (LAS)architectures, in accordance with various embodiments of the presentdisclosure.

FIGS. 3A-3D, 4A-4B and 5A-5B illustrate characteristics of the LASarchitectures of FIGS. 2A and 2B, in accordance with various embodimentsof the present disclosure.

FIG. 6 illustrates an example of a LAS architecture, in accordance withvarious embodiments of the present disclosure.

FIGS. 7A and 7B illustrate examples of lens and antenna arrangementsthat can be used in the LAS architectures of FIGS. 2A, 2B and 6, inaccordance with various embodiments of the present disclosure.

FIGS. 7C-7D and 8A-8B illustrate performance characteristics of the lensand antenna arrangements of FIGS. 7A and 7B, in accordance with variousembodiments of the present disclosure.

FIGS. 9A and 10A-10B illustrate an example of an antenna array using thelens and antenna arrangements of FIGS. 7A and 7B, in accordance withvarious embodiments of the present disclosure.

FIGS. 9B-9C and 10C-10F illustrate performance characteristics of theantenna array of FIGS. 9A and 10A-10B, in accordance with variousembodiments of the present disclosure.

DETAILED DESCRIPTION

Disclosed herein are various examples related to hybridmultiple-input/multiple-output (MIMO) architectures. Beam steering canbe provided using lens arrays. The data-rate impact of sucharchitectures can be considered from the MIMO signal processingperspective to optimize the system performance. One feature ofmillimeter (mm)-wave communications is high-gain antenna arrays forovercoming the attenuation and blockage challenges. Unlike the lowerfrequency bands, cost and power consumption of analog-to-digital anddigital-to-analog (ADC/DAC) components prohibit realizing mm-wave MIMOarchitectures fully in the baseband. Reference will now be made indetail to the description of the embodiments as illustrated in thedrawings, wherein like reference numbers indicate like parts throughoutthe several views.

Analog beamforming is a natural choice for alleviating the challenges ofaccommodating power hungry RF chains and ADC/DAC for each antennaelement. FIG. 1A is a schematic diagram illustrating an example of atraditional phased array (TA) architecture with variable phase shifters103 that can be used for analog beamforming. In this architecture,beamforming is accomplished by including a phase shifter (PS) 103 perantenna element 106. Phase shifters 103 are typically followed bytransmit/receive A&S (amplification and switch) stages 109 to meetradiated and/or received power requirements. FIG. 1B is a schematicdiagram illustrating an example of the Tx/Rx A&S 109 comprisingsingle-pole double-throw (SPDT) switches for coupling a power amplifier(PA) or low noise amplifier (LNA). Due to the significant reduction inradio frequency (RF) chain and ADC/DAC 112 number (e.g., reduction downto one as shown in FIG. 1A) the power consumed by the phase shifters 103(and their variable gain amplifiers) of large format traditional arraysbecomes an important power efficiency consideration. Larger antennaarrays are also expected to need phase shifters 103 with more bit statesand lower quantization errors to achieve desired beamformingperformance, further exacerbating the power efficiency issue. Mostimportantly, the data rate of the traditional phased array architectureis ultimately limited due to the inability to support simultaneousmultiple MIMO stream transmissions.

The desire to provide multiple MIMO stream transmissions whilemaintaining a reduced number of RF chains and ADCs/DACs has led to theintroduction of hybrid MIMO architectures. The usual hybrid MIMOarchitecture is based on the traditional phased array approach as shownin FIG. 10. Here, the RF signal from each RF chain 112 (a total numberof L) is split into antenna elements 106 (a total number of N) andpassed through phase shifters 103 before the antenna interface. Hence,the total number of phase shifters 103 (N_(PS)=NL) and power consumptiondramatically increase. This usual hybrid MIMO architecture isessentially a superposition of L phased arrays with each RF chain havingthe capability to form and steer the narrowest possible beam with theaid of its own phase shifter network (i.e., RF precoder/combiner).

More recently, the high power consumption and complexity of hybrid MIMOarchitectures motivated the consideration of alternative architecturesthat exhibit reduced number of phase shifters or replace phase shifterswith more energy efficient switches. The spectral efficiencies of thesealternative architectures have been evaluated as a function of thenumber of RF chains by introducing suitable MIMO channel estimationalgorithms. In addition, a power consumption model relying on expecteddissipated power in hardware components has been introduced tocharacterize the efficiency trade-offs among these alternativearchitectures. Spectral efficiency of the usual hybrid MIMO architecturehas been found to be significantly superior, thereby achieving capacityof the channel with the fewest number of RF chains. However, for equalpower consumption, all considered alternative architectures and theusual hybrid MIMO architecture have been found to perform with closespectral efficiencies. This is due to the fact that the alternativearchitectures consumed less power per RF chain, offered less capacityper chain, but captured the channel capacity with increasing number ofRF chains. Larger number of RF chains (in comparison to the usual hybridMIMO) in turn increased the power consumption and negated the powerefficiency benefit offered by the architecture.

Investigation of the alternative hybrid MIMO architectures has shownthat the hardware variations prohibit the capability of RF chains toform and steer the narrowest possible beam since phase shifters existonly for a subset of antennas or are removed and/or replaced in favor ofswitches. Hence, these alternative architectures result in a reducedspectral efficiency per RF chain. This disclosure introduces a novelhybrid MIMO architecture that addresses the outstanding challenges ofmm-wave networks: a superior spectrum efficiency under equal powerconsumption and cost effective hardware integration. Specifically, alens antenna subarray (LAS) architecture is proposed to strategicallyreduce the number of phase shifters and introduce energy efficientswitches.

FIG. 2A is a schematic diagram illustrating an example of a millimeter(mm)-wave lens antenna subarray (LAS) architecture with variable phaseshifters and SP(M)T (single-pole multi-throw) switch networks. Thearchitecture offers significant flexibility for making trade-offsbetween the total number of switches 215 and phase shifters 203 byadjusting the number of antennas 206 placed under each lens 218 (M).Each lens 218 is essentially a sub focal plane antenna array within theentire array composition: switching on an antenna generates a high gainbeam towards a particular direction. The beams that can be generated bythe antennas 206 under a lens 218 overlap to provide a continuousbeam-steering field of view (FoV).

For the same antenna aperture size, the LAS provides significantreductions in total phase shifter 203 count by resorting to antennaselect switches 215 while providing the electronic beam-steering andbeamwidth (i.e., gain) performance. LAS hardware configuration dependson the number of antennas per lens (M). The arraying of lens antennasubarrays and properly adjusting the phase of radiation via phaseshifters 209 provides array gain (in addition to lens gain) and cancapture the identical beamwidth performance that can be attained fromthe traditional phased array. Hence, LAS offers electronicbeam-steering, array gain, and narrow beamwidth performance like theusual hybrid MIMO architecture while utilizing significantly fewer phaseshifters to achieve a low power consumption, low-cost, and small areacircuitry. FIG. 2A depicts a scenario with an antennas per lens ratio ofM=4. LAS and TA hardware needs are directly proportional to the RF chainnumber (L). FIG. 2B is a schematic diagram illustrating an example of ahybrid MIMO architecture with the mm-wave lens antenna subarray (LAS)architecture. The RF signal from each RF chain 212 (1, 2, . . . L) issplit into antenna elements 206 (a total number of N) and passed throughphase shifters 203 and antenna select switches 215.

Note that the classical lens antenna array (LAA) approach uses a singlelens with no phase shifters but an N-way switch. Although this seemsadvantageous due to the removal of all phase shifters, the LAA techniqueis impractical for multiple reasons: a) for large array sizes, a verylarge lens is needed and the focal plane is deep into the lens apertureimplying a high profile assembly; b) all radiated power needs to passthrough a single antenna which stretches the demand on power amplifiersand power handling of Tx/Rx switches in the A&S blocks. The LASarchitecture allows adjustment of radiated power per lens and can leadto further RF hardware simplification by reducing the number of A&Sblocks. A hardware simplification is illustrated in FIG. 2A where, e.g.,the Tx/Rx A&S 209 coupled to the antennas 206 for the leftmost lens 218can be consolidated into a single Tx/Rx A&S 209 coupled between thephase shifter 203 and the switching network 215. For mm-wave MIMOarchitectures, the LAS based hybrid MIMO architecture offers lower pore,higher spectral efficiency and security. For equal power consumption,the LAS based hybrid MIMO can achieve higher spectral efficiency. Newestimation and system algorithms can be used for LAS. For mm-wave arraysin hybrid MIMO with an N element array and L RF chains, an M element perlens architecture results in (N*L)/M phase shifters and SP(M)T switchnetworks which can provide significant savings in power (e.g., >3×),smaller IC area, and lower costs.

Many of the existing MIMO algorithms do not jointly consider hardwarerequirements and limitations in the RF domain. However, alternative MIMOarchitectures have been investigated in terms of hardware limitations byconsidering power efficiency criteria. Unfortunately, these have failedto achieve increases in spectral efficiency under identical powerconsumption. In addition, the MIMO architecture investigations often donot pay sufficient attention to the importance of theintegration/assembly/test studies. Therefore, other realistic andpotentially serious issues (e.g., beyond the power consumption ofindividual IC units) continue to remain unaccounted for. For example,the usual hybrid MIMO architecture for N antennas utilizes multiple ofN:1 power splitting/combining hardware in the RF domain, which can bequite a significant challenge by itself for protecting the overallhardware power efficiency.

The proposed LAS architecture offers a solution to these issues, sincepower splitting is reduced by the number of lens elements and the switchnetworks effectively take care of the remaining splitting/combiningfunctionality. Similar to this example, a major contribution of thisdisclosure is to establish a superior but practical mm-wave network byjointly considering, optimizing, and/or linking the system and hardwarethrough a unification of communication, RF hardware, IC and packagingperspectives. This can result in low-cost, power-efficient, andspectrally-efficient mm-wave networks.

This disclosure will detail designs with volumetric lenses above theantenna elements. However, there are other RF circuits/networks that canbe connected to antennas to achieve beam-steering with switches withoutneeding phase shifters. These networks (usually referred to asRF/microwave beamforming networks) can also be placed below andconnected to the antennas to achieve a lower profile design andimplement the LAS based Hybrid MIMO architecture. Examples of suchRF/microwave beamforming networks include Rotman lenses and Butlermatrices.

Power consumption in traditional hybrid MIMO and alternative hybrid MIMOarchitectures has been investigated by considering the number of totalhardware components (e.g. phase shifters, switches) and typical powerconsumption expected from these components. A wide range of powerconsumption values can be observed and power consumption of particularhardware can be selected based on certain justifications such as averagevalue, expected trend in future years, etc. Here, a similar approach wastaken at the component level to evaluate the power consumption trends ofproposed lens antenna subarray (LAS) based and traditional phased array(TA) based architectures. First, these architectures were compared byassuming a single RF chain and ADC/DAC, e.g., using the architectures inFIGS. 1A and 2A with L=1. The desired effective isotropic radiated power(EIRP) were set to 45 dBm based on existing/expected standards. The peakbroadside gain of a traditional linear patch antenna array (TAG) can becalculated by assuming a 90% aperture efficiency (η_(TA)) asTAG=(4π/λ²)A_(p)η_(TA), where A_(p) stands for the footprint area. TheA_(p) is linearly proportional to the element number N and elementspacing d that is taken as d=λ/2 with λ representing wavelength.

The aperture efficiency of extended hemispherical or slab dielectriclenses is as high as patch antenna arrays when compared with respect tothe “footprint of lens base (=A_(p))”. Due to the lens/air mismatch andattractiveness of using low-cost (but with slightly higher loss)materials such as ABS, aperture efficiency of LAS (η_(LAS)) can be setto 80% and gain was evaluated as LASG=(4π/λ²)A_(p)η_(LAS). In the caseof cylindrical dielectric slab lenses, A_(p) of LAS and TA are equal asthe arrays have the same physical footprint. Total transmit power cantherefore be evaluated as TxP_(TA)=EIRP−TAG(dB) andTxP_(LAS)=EIRP−LASG(dB). FIG. 3A depicts examples of array gains andtransmit powers needed for both architectures (Tx power for 45 dBm EIRPand gain of TA and LAS) as a function of N. Array elements N=20corresponds to 16 dBi gain and agrees with the characterized gain of thepreliminary prototype shown in previous section. For N=64, the gain foreach approach is about 23 dBi and the needed power is about 23 dBm.

FIG. 3B depicts examples of the total number of hardware components(switches and phase shifters) needed by the two architectures (TA andvarious LAS implementations with L=1 RF chain). For the TA, the switchcount is assumed to be zero and all signal routing is achieved throughpower divider/combiner networks. The LAS can be implemented in variousways. Consider the possibility of using SP2T, SP4T, and SP8T switchesfor M antenna per lens cases of 4, 8, and 16. Assume a switch insertionloss of IL_(SP(M)T)=1 dB, which is a conservative value expected fromthe IC process utilized. The IL introduced by the SP2T implementationsis higher due to the need for using multiple switches in series. Forexample, SP4T and SP8T switch functionalities can be implemented with atotal of 3 and 7 SP2T switches and IL of 2 dB and 3 dB, respectively.FIG. 3B demonstrates that the numbers of phase shifters and switchesdecrease with larger M as expected. As compared to the traditional array(TA), a significant reduction is achieved in total phase shifters. ForN=64, the TA utilizes 64 phase shifters while the M=4 LAS uses 16 phaseshifters.

The power consumption of the TA architecture can be estimated asP_(TA)=TxP_(TA)/η_(PAPS)+NP_(PS)+P_(RFC), where η_(PAPS) stands for theefficiency of the transmit amplifier and phase shifter in series, P_(PS)stands for power consumption of phase shifters, P_(RFC) stands for thepower consumed in the RF chains and ADC/DAC. LAS is expected to providefurther advantage in the IL of N-way power dividers since switches alsoperform power division. The efficiency η_(PAPS) is independent of thenumber of Tx/Rx A&S blocks and phase shifters, as the total transmitpower is divided and passed through a single block. Phase shifters arein general implemented as active or passive devices. Although passivephase shifters in theory can provide negligible power consumption, inpractice they are combined with a variable gain amplifier (VAG) tobalance the IL variation among their phase shifting states. Hence,active or passive phase shifters contribute to the power consumptionsignificantly in very large arrays and in hybrid MIMO architectures.

FIG. 3C illustrates examples of power consumption in TA and LASarchitectures with L=1 RF chain. The curves shown in FIG. 3C aregenerated with η_(PAPS)=0.2, P_(PS)=100 mW, and P_(RFC)=500 mW. TheP_(RFC) was taken as significantly higher than P_(PS). The P_(PS) wasbased on commercially available phase shifters. The P_(sp2T) was takensignificantly lower as P_(SP2T)=10 mW. For power consumption of largerthrow switches, P_(SP4T)=20 mW and P_(SP8T)=40 mW were used. The powerconsumption of the LAS implementation with M antennas per lens wascalculated asP_(LAs)=TxP_(LAS)/(η_(PAPS)η_(SP(M)T))+(N/M)P_(PS)+N_(SP(M)T)P_(SP(M)T)+P_(RFC),where N_(SP(M)T) represents the total number of SP(M)T switches (orswitch networks) and η_(SP(M)T) stands for the efficiency of theswitches. This is related to the switch IL as η_(SP(M)T)=10^((−mIL)^(SP(M)T) ^(/10)) with m representing the number of series switchesneeded to implement the architecture. FIG. 3C demonstrates that forsmall arrays with N<30, the efficiency of amplifiers and phase shiftersdominates the power consumption. However, for larger arrays, powerconsumption in control components is the dominant contributor and largerlenses with more multiple throw switches offer better power savings.FIG. 3D illustrates an example of the Tx power needed to be generatedunder each lens within LAS architecture. FIG. 3D demonstrates that theRF power per lens also increases with lens size, stretching the A&Srequirements and potentially prohibiting the use of one A&S per lens.Hence, lens size must be strategically selected to provide the besthardware implementation in terms of efficiency and cost.

Take-aways from the foregoing analysis are that power consumptionreductions >30% are possible for N>28 (FIG. 3C) and lower P_(PS) valuesdo not alter the fact that significant power savings are achieved by LASin large array settings. For example, P_(PS)=50 mW achieves >30% powerconsumption reduction for N>40. The power savings achieved by the LASarchitecture further increases with the number of RF chains. Powerconsumption in usual hybrid MIMO with L chains can be modeled asP_(TA(L))=TxP_(TA)/η_(PAPS)+LNP_(PS)+LP_(RFC). Power consumption in LASbased architecture can be modeled asP_(LAS(L))=TxP_(LAS)/(η_(PAPS)η_(SP(M)T))+L(N/M)P_(PS)+LN_(SP(M)T)P_(SP(M)T)+LP_(RFC).

FIGS. 4A and 4B demonstrate examples of the hardware count and powerconsumption of L=4 and L=8 hybrid MIMO architectures. FIG. 4A showstotal number of components needed by traditional array (TA) and variousLAS implementations with L=4 and L=8 RF chains and FIG. 4B shows thepower consumption in TA and LAS architectures with L=4 and L=8 RFchains. LAS architectures were selected to exhibit identical total lensand RF chain numbers. To demonstrate fewer data points with moreclarity, the total number of lenses in LAS was set to be equal to thenumber of RF chains (i.e. N/M=L). FIG. 4B depicts that the traditionalhybrid MIMO power consumption becomes heavily dominated by the phaseshifters with increasing L; for N=64 the power for L=4 and L=8 TAdesigns is 30 W and 56 W, respectively. Significant power savings can beachieved by the LAS architecture even in small array settings. Forexample, the power consumption ratio between traditional array and LASbased hybrid MIMO is >2:1 for N=32 and L=8 when LAS is implemented withM=4 elements per lens, N/M=8 lenses and SP4T switches.

The proposed LAS hybrid MIMO architecture provides the flexibility torealize large antenna arrays with various lens sizes. For M=N, the LASarchitecture employs only a single lens. For small values of M (i.e.M<<N), LAS architecture gets closer to the traditional phased array.Hybrid MIMO systems that have control over individual antenna elementsforming the radiating aperture can take advantage of analog precodingand/or combining to align angularly spread multiple channel paths toincrease power gain. This is in contrast to the beamforming approachthat focuses beams only towards the best channel path that maximizespower. Hence, the LAS architecture operated with analogprecoding/combining naturally outperforms the beamforming approach basedon a single lens. It is important to identify a LAS architecture withoptimum M value based on joint investigations on spectral efficiency,power consumption, and hardware implementation complexity (i.e power perlens). The case of M=1 implies a traditional array, but is inefficientin power consumption.

As an example, consider an N=64 element LAS architecture with M=4, 8, 16and 64 to demonstrate that multiple focused beams offered by smalllenses still offer a competitive (within 10%) spectral efficiency withrespect to the traditional antenna array (M=1) when their beams areproperly precoded and combined. For demonstration, a NYU channelsimulator was used under non-line-of-sight (NLOS) 28 GHz scenario togenerate random wireless channels. Users were randomly distributedbetween 60 m to 200 m from the transmission point and the channel forthe users were provided by the simulator. Antenna capabilities of thearchitectures (e.g., gains, beamwidths, and array factors) and channeldispersion characteristics (e.g., angular power dispersion, multipathclusters, multipath phases, and path loss) were also modeled. Using thischannel information, precoding matrices were calculated and receivedpower by the users were determined along with their data rates.

FIGS. 5A and 5B, shown are examples of user capacity distribution in themodeled NLOS mm-wave environment using a traditional array (TA) andvarious LAS implementations with M=4, 8 and 16; and corresponding energyefficiency distribution, respectively. LAS architectures outperform theTA by more than 2:1 with some configurations being better than others.FIG. 5A illustrates the cumulative distribution (CDF) of the capacity ofthe users for 2 GHz bandwidth with 45 dBm EIRP (in accordance withprevious section's power consumption studies). It is observed that thetraditional array offers the best capacity in NLOS as expected. LASarchitectures with smaller lenses (M=4 and M=8) almost offer the samespectral efficiency by being within 10% of that capacity. However,larger lenses suffer in these NLOS situations. As such, the single lens(M=64) offers only 67% of the TA capacity. The benefit of the LASarchitecture is clearly observed when energy efficiency (i.e., data rateper Watt) is considered. FIG. 5B was obtained by dividing data rates ofFIG. 5A with power consumption evaluated in FIG. 3C. All LASarchitectures are capable of providing better spectrum efficiency underequal power consumption. Due to the variations in hardwareimplementation and data rates, some LAS architectures seem to be optimum(e.g., M=8 with 3.6 Gbps/W). Significant improvement in energyefficiency may be achieved by replacing TA (1.45 Gbps/W) with LAS.

As previously discussed, the power consumed by the PSs 103 (and theiramplifiers) can become prohibitive in large antenna arrays. This problemis exacerbated in traditional hybrid MIMO architectures (FIG. 1A) sincethe number of PSs 103 becomes a multiple of the number of ADC/DACsemployed within the architecture. Besides these recent considerations,the PS related hardware complexity has long been recognized as achallenge. For example, switched beam antennas utilizing beamformingnetworks (BFNs) are found attractive due to their PS-free nature;however, these suffer from large sizes. Likewise, TAs consisting ofsubarrays use significantly fewer PSs 103, but steer their beams withina narrow field of view (FoV). Significant PS 103 number reduction may beachieved by introducing variable gain amplifiers to subarrays; however,this approach still exhibits a narrowed FoV and is likely to suffer frompower consumption issues in large array settings.

Microwave lenses can be utilized to realize PS-free switched beamantennas, but they can be unattractive due to their size andhigh-profile focal surface array assembly. For beam steering arrays,transmit and receive (Tx/Rx) amplifier and switch (A&S) placement isalso a consideration. In TAs and BFN/lens-based switched beam antennas,the PS losses and switch power handling capabilities usually need theA&S blocks to be placed behind each antenna element.

A novel mm-wave beam steering antenna comprising lens antenna subarrays(LASs) is introduced. FIG. 6 shows an example of the LAS-based antennawith Tx/Rx A&S 209 positioned between the PS 203 and SP(M)T switchnetworks 215. As depicted in FIG. 6, each LAS hosts M=5 antennas to forma switched beam array. A total of L LASs are interfaced with L PSs torealize a similar beamwidth performance with a TA having a similarnumber of antenna elements (N≈LM). The choice of M allows tradeoffsamong the switch network throws, PS numbers, lens size, and power perlens, thereby reducing the number of PSs 203 and potentially A&S blocks209 as well. The combination of beam switching and steering via PSs hasattracted interest for increasing FoV and providing higher steeringresolution without being concerned with the number of PSs. Thecombination has been towards reduced PS numbers; however, a narrow FoVas in traditional subarrays is achieved due to restricting the antennasof the subarrays with 1 bit PSs. The LAS-based antenna presented in thisdisclosure offers an improvement by reducing the PS numbers whilesimultaneously maintaining a large FoV that is determined by the lensand its focal surface type (e.g., curved versus planar). An example of aLAS-based antenna at 38 GHz is demonstrated here.

Lens and Feed Antenna Design

For demonstration, a LAS-based antenna design was carried out for ascenario of N=20, M=5, and L=4. Based on the full-wave simulation of anN=20 element patch antenna based TA and aperture efficiencyconsiderations, obtaining >15 dBi broadside realized gain with theLAS-based antenna is important to demonstrating a comparable radiationperformance. The LAS-based antenna architecture in FIG. 6 implies thatthe FoV and bandwidth performances primarily depend on the lens and feedantennas, respectively. To utilize three-dimensional (3-D) printing,extended hemi-cylindrical dielectric slab waveguide (DSW) lenses fromthermoplastic acrylonitrile butadiene styrene (ABS, ϵ_(r)=2.6, tanδ=0.006) are selected for the design. The lens was fed by a 38 GHzaperture-coupled patch antenna exhibiting 7% |S₁₁|<−10 dB bandwidth tobe suitable for operation within the 37 and 39 GHz mm-wave bands. Thepatch antenna was designed to be separated from the lens by an air gapof 0.25 mm. The air gap enables mechanically moving the lens over thefeed antenna and demonstrates the concept without implementing theswitch network.

The 50Ω microstrip feedline, ground plane, coupling aperture, and patchantenna were designed within a multilayered printed circuit board (PCB)substrate stack-up shown in FIG. 7A, which provides the lens dimensionsand substrate stack-up for the feed antenna. The substrate materialswere Rogers RO4003 with ϵ_(r)=3.35 and tan δ=0.027 (see FIG. 7A for thesubstrate thicknesses). In this disclosure, Ansys HFSS has been utilizedas the full-wave electromagnetic simulation tool. A patch antenna designwas carried out by representing the presence of the lens above theantenna with a 2 mm thick rectangular prism and enclosing the entiresolution domain with radiation boundary conditions. The global materialenvironment was set to be the ABS to model an infinitely large lensabove the antenna. The designed antenna layout was found to operate wellwith the finite lens as well. FIG. 7B depicts the layout of the patchantenna and substrate footprint dimensions (in mm) used with theinfinitely large lens model.

In the lens design, the E-plane (xz plane) of the patch antenna isaligned with the width of the lens to excite the symmetric TM₀ mode. Thewidth of the lens was selected as 5 mm to be relatively large ascompared to the patch antenna footprint. The lens diameter is determinedas M(λ₀/2)≈20 mm, where λ₀=7.9 mm stands for the free-space wavelengthat 38 GHz. For the selected material and thickness, the solution oftranscendental equations arising from the DSW problem at 38 GHz revealseffective relative dielectric constant of the TM₀ mode as ϵ_(re)=2.06.An initial value for the extension length L_(Ex) is determined from theequations in “Double-slot antennas on extended hemispherical andelliptical silicon dielectric lenses” by D. F. Filipovic et al. (IEEETrans, Microw. Theory Techn., vol. 41, no. 10, pp. 1738-1749, October1993) as L_(Ex)=13.65 mm.

A cylindrical surface was adapted (instead of an elliptical surface) toachieve a larger FoV. For volumetric lenses, it has been demonstratedthat full-wave simulations for the design and modeling of small lensesis needed. This disclosure appears to be the first to report arelatively small extended DSW lens. Therefore, the full-wave simulationswere carried out to identify the L_(Ex) value that maximizes therealized gain with substrate footprint of the feed antenna taken equalto that of the lens base. The parametric sweep for L_(Ex) was carriedout in 1 mm increments within a 13.5-20.5 mm range to keep a low antennaprofile. FIG. 70 shows an example of the simulated |S₁₁| of the feedantenna within the 34-42 GHz band. As depicted in FIG. 70, the feedantenna remains impedance-matched for all L_(Ex) values. It is observedthat |S₁₁| exhibits more ripples for larger L_(EX) values, indicatingmore reflections within the lens that could potentially contribute toincreased sidelobe levels. FIG. 7D presents the simulated realized gainas a function of the extension length L_(Ex) of the lens at 38 GHz.L_(EX)=15.5 mm is selected due to its maximum gain value of 13.5 dBi.The simulated radiation efficiency is 78%.

LAS-Based Antenna Performance

To evaluate the antenna, the beam steering performance of a single lenswhen its feed antenna is located at different positions (my) relative toits center axis was evaluated. Although the mechanical movement employedin the evaluated design allows positioning the feed antenna anywherewithin the focal surface, determining the fewest number of antennalocations that can cover the FoV is very advantageous forswitch-network-based array implementations with the lowest possiblehardware complexity. The xy-plane (E-plane, φ=0°) and yz-plane (H-plane,φ=90°) simulated realized gain patterns of the feed antenna located inthe middle of the focal surface (my=0) are shown in FIG. 8A. Thesimulated realized gain patterns are from a single lens at 38 GHz whenthe feed antenna is located at different positions (my). Due to the slabgeometry of the lens, the E-plane gain pattern exhibits a 38° half-powerbeamwidth (HPBW). In contrast, the H-plane gain pattern is well focusedwith a 15° HPBW. The sidelobes are below 12 dB of the major beam, andtheir characteristics are related to the lens type and size.

Back radiation can further be minimized by utilizing cavity-backedapertures; however, this is not pursued in the presented design. Basedon the 15° HPBW, the feed antenna positions (my) corresponding tobeam-steering directions of 15°, 30° are of interest to generate beamswith HPBW overlap and have the fewest possible antennas in a switchednetwork implementation. Parametric simulation studies are utilized toidentify these positions as my=−8, −4, 0, 4, and 8 mm. FIG. 8A alsodepicts the realized gain patterns for my=4 and 8 mm. Specifically,my=±4 mm and my=±8 mm result in peak realized gains of 12.45 and 11.86dBi, respectively. Larger my values do not provide significant beamsteering capability and provide increased sidelobes due to being closeto the lens boundary and out of focus. The FoV of the lens can,therefore, be defined as 75° from the maximum beam-steering angle andthe HPBW consideration.

The beam steering performance of the designed antenna with L 4 can bepredicted from an array theory by multiplying the single-lens gainpattern with the array factor (AF). As expected, the electrically largespacing between lenses causes the AF to exhibit grating lobes. FIG. 8Billustrates an example of the AF pattern at 38 GHz for the L=4 elementLAS when the AF is steered towards −5°, 0°, and 5° directions. Negativemy values generate symmetric patterns with respect to 0°, which are notshown. FIG. 8B depicts the normalized AF patterns of the L=4 antennawith 20 mm lens spacing in positive half of the H-plane when the AF issteered within the HPBW of the feed antenna located at the center of thefocal surface (my=0). Since the grating lobes are out of the major beam,the high-gain nature of the antenna remains intact. Depending on theoverlap between the grating lobes and sidelobes of the lens, sidelobeswith narrower beamwidth are also expected. The AF exhibits 5.2° HPBW,The AF steered in 5° increments therefore results in patternsintersecting each other approximately about their HPBW. Hence, the beamsteering performance of the designed antenna is depicted withrepresentative patterns steered in 5° increments. The AF can be steeredin much finer resolution based on the PS capabilities included in theLAS-based antenna architecture.

The placement of multiple lenses adjacent to each other necessitatesutilizing full-wave simulations for accurately evaluating the radiationperformance of the LAS-based antenna. FIG. 9A is a graphicalillustration of a full-wave simulation model of the 4-element LAS-basedantenna. Representative simulated realized gain patterns at 38 GHz fordifferent my positions are shown in FIGS. 9B and 9C without and withconductive walls, respectively. Negative my values generate symmetricpatterns with respect to 0°, which are not shown. As shown in FIG. 9A,the lenses 218 are in contact to utilize the smallest possible spacing(e.g., 20 mm), To accommodate the placement of 2.92 mm edge connectorsfor the experimental setup, the microstrip line substrate was extendedby 5 mm from the edge of the 15 mm wide substrate stack-up that includesthe feed antennas. The holes in the substrates were for mounting theedge connectors and a 3-D printed holder. Simulations were performed formy=0, 4, and 8 mm positions. The excitation phases were calculated torealize AF patterns with major beams within the single-lens beamsteering cone (e.g., −7.5° to 7.5° for my=0, 7.5° to 22.5° for my=4 mm,and 22.5° to 37.5° for my=8 mm). The excitation phases were also roundedbased on a 5-bit phase quantization. FIG. 9B presents the representativeH-plane realized gain patterns steered with 5° increments. Maximum andminimum realized gains are 16.37 and 13.4 dBi for the 0° and 35°directions, respectively. The simulated radiation efficiency was 71% andslightly lower than that of the single lens. The loss from the extendedfeedline is a major contributor of this deviation. Within the scanrange, the sidelobes are below 8.6 dBi. The worst-case simulated mutualcoupling was −27.5 dB at 37.4 GHz.

For applications such as point-to-point communications, it is generallydesirable to have sidelobe levels that are at or below 10 dB of the mainbeam. For THz focal plane arrays, one potential effect that distorts thegain pattern is the illumination of a lens surface by the feed antennasof the adjacent lenses. To investigate the possibility of reducing thesidelobe level, the vertical conductive walls have also been consideredas shown in FIG. 9A. The corresponding simulated realized gain patternsare shown in FIGS. 9B and 90. It can be observed that the sidelobes aresuppressed more in a way to approach to the 10 dB level. Specificallywith walls, the maximum and minimum realized gain values were 17 and15.41 dBi for the 0° and 20° directions, respectively. Wthin the scanrange, the sidelobes are below 7.0 dBi.

Experimental Verification

FIG. 10A is an image depicting the antenna assembly used forverification, and FIG. 10B is an image of the antenna prototype (withoutconductive walls). Mechanical movement is used to demonstrate theantenna performance without the SPST network implementation. Amicroservo motor with 3.2×1.2×2.9 cm3 volume, 9 g weight, and 1 W dcpower consumption (<0.05 W in idle state) is interfaced with amicroprocessor to control the movement accuracy. The feed antennas wererealized with multilayered PCB fabrication and interfaced with 2.92 mmedge connectors. The lenses and the linear gear were fabricated as asingle part by making use of an nScrypt 3Dn printer. The antennaprototype does not include the conductive walls for fabrication ease.The gear and the customized holder were also fabricated with the same3-D printer. The motor and the parts were assembled together withplastic screws. The PS integration was emulated in software bymultiplying and/or summing the measured individual lens patterns andusing a 5-bit phase quantization. The total motion range required tosteer the beam within the FoV is 16 mm (my=8 mm). This can be traversedin 100 ms based on the selected motor model and gear size. FIG. 9Bdepicts the antenna prototype.

The measured |S₁₁| performance of the feed antennas is shown in FIG. 10C(measured reflection coefficients from the feed antennas). Specifically,the antennas are well matched with 8.5% |S₁₁|<10 dB bandwidth andexhibit slight center frequency shifts with respect to each other. Thesebandwidth and center frequency shifts are likely due to the smalldimensional variations resulting from fabrication when etching thecoupling apertures. On the other hand, the slight variations insubstrate dimensions and their electrical properties (along with thecoupling aperture variations) could have resulted in the differencebetween simulated and measured center frequency. Nevertheless, theantennas are still well matched at 38 GHz.

FIGS. 10D, 10E and 10F depict representative H-plane measured gainpatterns at 38 GHz for different my positions of: my=0, my=4 mm, andmy=8 mm, respectively. The maximum and minimum gains were 15.7 and 13.15dBi for the 0° and 35° directions, respectively. Considering that edgeconnectors exhibit 0.5-0.7 dB insertion loss within 30-40 GHz band, themeasured and simulated gain performances are in excellent agreement.From this agreement, it can be concluded that the radiation efficiencyof the LAS antenna is almost identical to the simulated value of 71%(which includes the additional loss of the extended feedline forexperimental purposes).

A mm-wave beam steering antenna based on the LASs has been introduced.

As compared to a traditional phased array, the LAS-based antenna offersa reduced hardware complexity by utilizing multiple-throw switchnetworks to significantly reduce the number of PSs. The lens diametercan potentially be selected small enough to achieve a low profile andreduce the number of Tx/Rx amplifier and switch networks. A 38 GHzantenna design comprising L=4 LASs was presented to demonstrate theconcept. Each LAS utilizes the DSW-based hemi-cylindrical lens with fivefeed antennas. For rapid prototyping, mechanical movement over asingle-feed antenna was utilized to emulate the presence of these fivefeed antennas. The prototype performed with 8.5% |S₁₁|<10 dB bandwidth,15.7 dBi gain, 5.2° H-plane HPBW, and 75° beam steering range.

It should be emphasized that the above-described embodiments of thepresent disclosure are merely possible examples of implementations setforth for a clear understanding of the principles of the disclosure.Many variations and modifications may be made to the above-describedembodiment(s) without departing substantially from the spirit andprinciples of the disclosure. All such modifications and variations areintended to be included herein within the scope of this disclosure andprotected by the following claims.

The term “substantially” is meant to permit deviations from thedescriptive term that don't negatively impact the intended purpose.Descriptive terms are implicitly understood to be modified by the wordsubstantially, even if the term is not explicitly modified by the wordsubstantially.

It should be noted that ratios, concentrations, amounts, and othernumerical data may be expressed herein in a range format. It is to beunderstood that such a range format is used for convenience and brevity,and thus, should be interpreted in a flexible manner to include not onlythe numerical values explicitly recited as the limits of the range, butalso to include all the individual numerical values or sub-rangesencompassed within that range as if each numerical value and sub-rangeis explicitly recited. To illustrate, a concentration range of “about0.1% to about 5%” should be interpreted to include not only theexplicitly recited concentration of about 0.1 wt % to about 5 wt %, butalso include individual concentrations (e.g., 1%, 2%, 3%, and 4%) andthe sub-ranges (e.g., 0.5%, 1.1%, 2.2%, 3.3%, and 4.4%) within theindicated range. The term “about” can include traditional roundingaccording to significant figures of numerical values. In addition, thephrase “about ‘x’ to ‘y’” includes “about ‘x’ to about ‘y’”.

Therefore, at least the following is claimed:
 1. A lens antenna subarray(LAS), comprising: a plurality of antenna elements configured toselectively receive a radio frequency (RF) transmission signal from RFprocessing circuitry, the RF transmission signal provided to a selectedantenna of the plurality of antenna elements via a switching network, acommon phase shifter, and an amplification and switch (A&S) stage; and alens extending across the plurality of antenna elements, the lensconfigured to steer a RF transmission generated by the selected antennaof the plurality of antenna elements in a defined direction, theselected antenna determined by the switching network.
 2. The LAS ofclaim 1, wherein the RF processing circuitry comprises a RF chain thatprovides the RF transmission signal to the common phase shifter fortransmission by the selected antenna.
 3. The LAS of claim 1, wherein theplurality of antenna elements consists of M antenna elements and theswitching network and the switching network comprises a single-poleM-throw (SP(M)T) switch configured to switch between the M antennaelements to direct the RF transmission signal to the selected antennafor transmission through the lens.
 4. The LAS of claim 1, wherein thelens is a dielectric slab waveguide (DSW) lens.